Headphone Amplifier


This audio power amplifier in principle is a "Current Feedback" design (opposite to most amplifiers that are Voltage Feedback designs), see figure 1 below. The input signal is fed to an input buffer with a voltage gain of one and output impedance R0.

The resistors RF and RI attenuate the output signal and make the feedback network. The difference between the input signal and the feedback signal results in the error current I. This is led to a current mirror and transformed to an equivalent voltage across the resistor RT, which normally has a high value. This voltage is fed to the output via an output buffer, which is a current amplifier (with a voltage gain of one). CT is a compensation capacitor, which provide for the stability of the amplifier (and the necessary phase margin). Amplifiers of the Current Feedback design are taken out to get amplifiers where the bandwidth is nearly independent of the voltage gain. Amplifiers of this type have a very large bandwidth and are very fast.


Schematic description

The amplifier, view the circuit diagram below, is fully symmetrical built up as a voltage amplifier followed by a current amplifier. The relation R25/R6 approximately gives the closed loop gain as 20 dB with the values shown.

The input stage consists of two JFETs in a symmetrical coupling instead of the usual (minimum four) bipolar transistors. If one extract the difference between the drain current for these complementary field effect transistors, in theory a non-distortion amplifier can be realised. In practice the two transistors will be different, causing the difference current to superimpose even harmonic components. As it is used bipolar transistors in a symmetric coupling to make the difference current, these will add additional odd harmonic components. The final result is an amplifier reminding of a valve amplifier: The distortion consists of mostly second harmonic components followed by third harmonic etc, in other words symmetric falling harmonic components.

The offset adjustment is made up by the potentiometer P5. The resistors R1 + R2 set the input impedance. The resistor R1 will together with the source output resistance and the input capacitance of the amplifier (about 60 pF) set the upper cut-off frequency. If the source has e.g. 600 ohms output impedance, the cut-off frequency is about 1.7 MHz.

The field effect transistors J3 (N-channel) and J4 (P-channel) withstand a maximum of respectively 40 V and 20 V to operate properly, therefore the power supply voltage V+ and V- should not exceed 20 V.

We are using ordinary current mirror, Q11/Q13 for the positive leg and Q12/Q14 for the negative leg. The voltage on the input of the amplifier is transformed to a proportional current in the JFETs. This current is compared to the feedback current (via R25). The resulting error current is transformed to a voltage across R15 and R16. This is the summing point before the current amplifier.

The open loop gain is given by the JFET's transconductance reduced with local feedback, multiplied with the double load on the gate of M21/M22. Without the resistors R15 and R16 the load at this point is mostly capacitive and parameter dependent. Thus these resistors determine the open loop gain to about 45 dB. With a closed loop gain at about 20 dB, the feedback factor is relatively low: about 25 dB.

It is not added any capacitors to stabilize the amplifier, since the internal capacitances of the MOSFETs are quite high. If any instability should be a problem, try to insert two capacitors of about 47 pF each in parallell with R15 and R16. Without any compensation capacitors the Slew Rate limitation do not set in before over 50 V/µs.

The amplifier has the same open loop gain and bandwidth for all audio frequencies, since the open loop bandwidth is very high, about 60 kHz. This is equivalent with nearly equal amount of distortion and the same output impedance over the whole audible range. The output impedance is nearly resistive througout the whole audio range, only barely inductive at the highest frequencies (a few degrees at 20 kHz).

The phase margin is about 70 degrees at the given closed loop gain. If higher phase margin is desired, the capacitors mentioned above, may be inserted. These should not have any negative influence on the amplifier performance. The linearity is good, thanks to the use of high local feedback in the input stage in combination with the current mirror.

The current amplifier used is a symmetrical MOSFET source follower, chosen linearity requirements, and at the same time having good thermal stability. The bias generator consists of the components 17-20. The quiescent current is set by means of the potentiometer P19.

The regulated power supply is low pass filtered by means of R27 and R28 plus C31 and C32. Together with the decoupling capacitors C29 and C30 and the feedback, the low pass filtering ensures that there is no ripple and noise at the amplifier output. Adjustable voltage regulators U41 and U42 are used to deliver a regulated positive and negative 18 V supply, see the schematic below.

This value is set by the resistors R37/R39 (and R38/R40). The transformer should be connected to the bridge rectifier D52. A suitable transformer is a 2x18 V 30 VA toroid for both channels (or two separate transformers, if wanted).

The impedance of headphones is varying over several decades, and most of them lie within the range of 32-1200 ohms. The lower the impedance, the lower voltage is needed, the higher the current, however. The simulation results are presented with 30 mA quiescent current (M21 and M22). The quiescent current may be increased above this value depending on the heat sinks used. Heatsinks with 16K/W can be suitable.

Some simulation results

Max output voltage:
Output impedance:
Frequency range:
Slew Rate:
THD at 5 V peak:
Rise/Fall time:
Input impedance:

11 V peak
< 1.5 ohm, 10 degrees (1Hz-10kHz)
DC-1.3 MHz
< ± 70 V/µs
< 0.005 % (300 ohm load, 1 kHz)
< 1 µs
20 dB (10x)
180 kohm||60 pF

The layout (pdf file) is shown for two channels, totally separated. The component placement below is also shown for two channels, totally separated.

The printed circuit board measures 100x96 mm.

The parts list applies for both the circuit board and the power supply.

Mounting description

1/2 W metal film oxide resistors with 1 % tolerance may be used. The JFETs 2SK170BL/2SJ74BL should be used for J3/J4. It should also be possible to use 2SK170GR/2SJ74GR. The couple 2SK147GR/2SJ72GR could also be used instead. These have higher internal capacitances and transconductance and are slightly more linear. In return the price is higher. A replacement for the couple 2SC1775/2SA872 used for Q12/Q14/Q11/Q13 is e.g. 2SC1815/2SA1015 or 2SC2240/2SA970. IRF620/IRF9620 is used for M20/M21/M22. These may be replaced by IRF610/IRF9610. For all replacements, be sure that the pinning is correct when mounting the transistors on the circuit board.

The MOSFETs M21 and M22 are running at a current of nominally 30 mA and should be mounted with a small heat sink. There should not be any quiescent point drifting when the temperature for the MOSFETs has stabilised.

The two parallel outputs from the transformer are carried to the rectifiers, see the figure below.

From each channel board, make a connection to a common ground on the chassis. To this point, also make a connection from the mid tap of the transformer. Normally one will use a volume control connected to the amplifier input.

Start-up and adjustment

When the power supply voltage is working as wanted, adjust the output-offset voltage by means of the potentiometer P5 to be close to 0 V DC. Also adjust the quiescent current to initially be at a minimum, and increase this slowly by means of the potentiometer P19. If possible, look at the output with an oscilloscope, there should not be anything but noise here if everything is OK. It should not be necessary to re-adjust either offset voltage or quiescent current.

About 1.2 V peak input voltage is required before the output clips. For lower headphone impedances, the voltage is much lower. If higher gain is wanted, R6 is reduced (and vice versa). Please note that the feedback resistor R25 should be unchanged. Note that none of the good properties of the amplifier, like bandwidth, distortion and Slew Rate, are deteriorated by moderate change of R6.

Known problems

The voltage regulators U41 (LM317) and U42 (LM337) may fail with too large capacitive load. This can be cured by mounting diodes between input and output of the regulators. For LM317 the cathode is coupled to the input and the anode to the output. For LM337 the cathode is coupled to the output and the anode to the input. Parasitic oscillations for M21/M22 are cured by inserting small resistors (47-200 ohm) in series with the transistors gate.

Please notice:
This project description is for non-commercial use, only. Using this document on a site and charging a fee for download is vialation of non-commercial use and prone to demand for payment. So, for commercial use, contact me for agreement of terms. This page, however, can be downloaded for own use, and linked to, not violating term of non-commercial use.


Knut Harald Nygaard